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FEATURES
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LTC3700 Constant Frequency Step-Down DC/DC Controller with LDO Regulator DESCRIPTIO
The LTC(R)3700 is a constant frequency current mode stepdown (buck) DC/DC controller with excellent AC and DC load and line regulation. The on-chip 150mA low dropout (LDO) linear regulator can be powered from the buck controller's input supply, its own independent input supply or the buck regulator's output. The buck controller incorporates an undervoltage lockout feature that shuts down the controller when the input voltage falls below 2.1V. The buck regulator provides a 2.5% output voltage accuracy. It consumes only 210A of quiescent current in normal operation with the LDO consuming an additional 50A. In shutdown, a mere 10A (combined) is consumed. For applications where efficiency is a prime consideration, the buck controller is configured for Burst Mode operation which enhances efficiency at low output current. To further maximize the life of a battery source, the external P-channel MOSFET is turned on continuously in dropout (100% duty cycle). High constant operating frequency of 550kHz allows the use of a small external inductor. The LDO is protected by both current limit and thermal shutdown circuits. The LTC3700 is available in a tiny 10-pin MSOP.
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Dual Output Regulator in Tiny 10-Pin MSOP High Efficiency: Up to 94% Wide VIN Range: 2.65V to 9.8V Constant Frequency 550kHz Operation 150mA LDO Regulator with Current Limit and Thermal Shutdown Protection High Output Currents Easily Achieved Burst Mode(R) Operation at Light Load Low Dropout: 100% Duty Cycle Current Mode Operation for Excellent Line and Load Transient Response 0.8V Reference Allows Low Output Voltages Low Quiescent Current: 260A Total Shutdown Mode Draws Only 10A Supply Current Common Power Good Output for Both Supplies
APPLICATIO S
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Notebook Computers Portable Instruments One or Two Li-Ion Battery-Powered Applications
, LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a registered trademark of Linear Technology Corporation.
TYPICAL APPLICATIO
VIN1 5V C1 10F 10V R1 0.068
VIN SENSE - PGATE
VIN2 LDO 169k VFB2 78.7k
C3 10F 10V
VIN2 3.3V 90 VOUT2 2.5V AT 150mA 86 82 78
EFFICIENCY (%)
VOUT1 1.8V AT 1A C2 47F 6V
L1 10H 100k
M1
C4 2.2F 16V
D1
LTC3700
+
80.6k VFB 10k ITH/RUN 220pF GND PGOOD C1, C3: TAIYO YUDEN EMK325BJ106MNT C2: SANYO POSCAP 6TPA47M C4: MURATA GRM42-6X7R225K016AL D1: MOTOROLA MBRM120T3 L1: COILTRONICS UP1B-100 M1: Si3443DV 3700 F01 R1: DALE 0.25W
Figure 1. High Efficiency 5V to 1.8V/1A Buck with 3.3V to 2.5V/150mA LDO
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Buck Efficiency vs Load Current
VOUT = 1.8V RSENSE = 0.068 VIN = 3.3V
74 70 66 62 58 54 50 1 VIN = 4.2V
VIN = 5V
10 100 LOAD CURRENT (mA)
1000
3700 F01a
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LTC3700
ABSOLUTE MAXIMUM RATINGS
(Note 1)
PACKAGE/ORDER INFORMATION
ORDER PART NUMBER
TOP VIEW VIN2 LDO VFB2 PGOOD GND 1 2 3 4 5 10 9 8 7 6 ITH/RUN VFB SENSE - VIN PGATE
Buck Input Supply Voltage (VIN) ................- 0.3V to 10V SENSE -, PGATE Voltages ............. - 0.3V to (VIN + 0.3V) VFB, ITH/RUN Voltages ..............................- 0.3V to 2.4V PGATE Peak Output Current (< 10s) ....................... 1A LDO Input Supply Voltage (VIN2) .................- 0.3V to 6V LDO, VFB2 Voltages ..................... - 0.3V to (VIN2 + 0.3V) PGOOD Voltage .........................................- 0.3V to 10V LDO Peak Output Current (< 10s) ..................... 500mA Storage Ambient Temperature Range ... - 65C to 150C Operating Temperature Range (Note 2) ... -40C to 85C Junction Temperature (Note 3) ............................. 150C Lead Temperature (Soldering, 10 sec).................. 300C
LTC3700EMS
MS PACKAGE 10-LEAD PLASTIC MSOP TJMAX = 150C, JA = 230C/ W
MS PART MARKING LTXN
Consult LTC Marketing for parts specified with wider operating temperature ranges.
The q denotes specifications that apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = VIN2 = 4.2V unless otherwise specified. (Note 2)
PARAMETER Buck DC/DC Controller Input DC Supply Current Normal Operation Sleep Mode Shutdown UVLO Undervoltage Lockout Threshold Shutdown Threshold (at ITH /RUN) Start-Up Current Source Regulated Feedback Voltage Output Voltage Line Regulation Output Voltage Load Regulation VFB Input Current Overvoltage Protect Threshold Overvoltage Protect Hysteresis Oscillator Frequency Gate Drive Rise Time Gate Drive Fall Time Peak Current Sense Voltage Peak Current Sense Voltage in Burst Mode VFB = 0.8V VFB = 0V CLOAD = 3000pF CLOAD = 3000pF (Note 6) 500 VITH /RUN = 0V (Note 5), 0C to 70C (Note 5), -40C to 85C 2.65V VIN 9.8V (Note 5) ITH /RUN Sinking 5A (Note 5) ITH /RUN Sourcing 5A (Note 5) (Note 5) Measured at VFB 0.820
q q
ELECTRICAL CHARACTERISTICS
CONDITIONS Typicals at VIN = 4.2V (Note 4) 2.65V VIN 9.8V 2.65V VIN 9.8V 2.65V VIN 9.8V, VITH /RUN = 0V VIN < UVLO Threshold VIN Falling VIN Rising
q q q
MIN
TYP
MAX
UNITS
210 200 10 10 1.90 2.00 0.15 0.25 0.780 0.770 2.10 2.20 0.30 0.5 0.800 0.800 0.1 4 4 10 0.860 20 550 110 40 40 120 30
340 330 30 30 2.60 2.65 0.45 0.85 0.820 0.830
mV/V mV/A mV/A 50 0.910 650 nA V mV kHz kHz ns ns mV mV
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A A A A V V V A V V
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LTC3700
The q denotes specifications that apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = VIN2 = 4.2V unless otherwise specified. (Note 2)
PARAMETER LDO Regulator VIN2 Input Voltage Input DC Supply Current Normal Operation with Buck Enabled Normal Operation with Buck Undervoltage Shutdown with Buck Enabled Shutdown with Buck Undervoltage Regulated Feedback Voltage Output Voltage Line Regulation With Buck Enabled With Buck Enabled With Buck Undervoltage Output Voltage Load Regulation VFB2 Input Current LDO Short-Circuit Current LDO Dropout Overtemperature Trip Point Overtemperature Hysteresis PGOOD Feedback Voltage PGOOD Threshold PGOOD High-to-Low PGOOD Low-to-High PGOOD On-Resistance (Note 8) VFB or VFB2 Falling VFB or VFB2 Rising VFB or VFB2 Rising VFB or VFB2 Falling VITH/RUN = 0V, VIN = VIN2 = 4.2V, VPGOOD = 100mV - 12 - 10 - 7.5 7.5 - 5.0 5.0 135 % % % % VLDO = 0V VIN2 = 3.3V, ILDO = 150mA VIN2 = 6V, ILDO = 150mA (Note 7) (Note 7) 150 Typicals at VIN2 = 4.2V 2.4V VIN2 6V 2.4V VIN2 6V 2.4V VIN2 6V, VITH/RUN = 0V 2.4V VIN2 6V, VITH/RUN = 0V 0C TA 70C, ILDO = 1mA -40C TA 85C, ILDO = 1mA (Unity-Gain Feedback) 2.65V VIN 9.8V 2.4V VIN2 6V, ILDO = 1mA 2.4V VIN2 6V, ILDO = 1mA 1mA ILOAD 150mA
q q
ELECTRICAL CHARACTERISTICS
CONDITIONS
MIN 2.4
TYP
MAX 6
UNITS V A A A A V V mV/V mV/V mV/V
50 100 0 8 0.780 0.765 0.800 0.800 0.05 4 4 0.06 0 200 270 170 150 5
100 150 1 24 0.830 0.835
0.12 10
mV/mA nA mA mV mV C C
12 10 180
Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: The LTC3700 is guaranteed to meet specifications from 0C to 70C. Specifications over the -40C to 85C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: TJ = TA + (PD * JAC/W)
Note 4: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. Note 5: The LTC3700 is tested in a feedback loop that servos VFB to the output of the error amplifier. Note 6: Peak current sense voltage is reduced dependent on duty cycle to a percentage of value as given in Figure 2. Note 7: Guaranteed by design; not tested in production. Note 8: PGOOD values are expressed as a percentage difference from the respective "Regulated Feedback Voltage" as given in the table.
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LTC3700 TYPICAL PERFOR A CE CHARACTERISTICS
BUCK DC/DC CONTROLLER
VFB Voltage vs Temperature
805 NORMALIZED FREQUENCY SHIFT (%) 804 803 VFB VOLTAGE (mV) 802 801 800 799 798 797 796 795 -55 -35 -15 5 25 45 65 85 105 125 TEMPERATURE (C)
3700 G01
VIN = 4.2V ITH/RUN = VFB NO LOAD
Shutdown Threshold vs Temperature
400 380 360 ITH/RUN VOLTAGE (mV) VIN = 4.2V
2.30 2.28 2.26
TRIP VOLTAGE (V)
340 320 300 280 260 240 220 200 -55 -35 -15 5 25 45 65 85 105 125 TEMPERATURE (C)
3700 G04
Buck Supply Current vs Input Voltage
250 240
VIN SUPPLY CURRENT (A)
230 220 210 200 190 180 170 160 150 2
ITH/RUN = VFB VIN2 = 0V TA = 25C
TRIP VOLTAGE (mV)
3
8 6 5 4 7 VIN INPUT VOLTAGE (V)
4
UW
Normalized Oscillator Frequency vs Temperature
10 8 6 4 2 0 -2 -4 -6 -8 -10 -55 -35 -15 5 25 45 65 85 105 125 TEMPERATURE (C)
3700 G02
VIN = 4.2V
Undervoltage Lockout Trip Voltage vs Temperature
2.24 2.20 2.00 2.18 2.16 2.14 2.12
VIN RISING
VIN FALLING
2.10 -55 -35 -15
5 25 45 65 85 105 125 TEMPERATURE (C)
3700 G03
Maximum (VIN - SENSE -) Voltage vs Duty Cycle
130 120 110 100 90 80 70 60 VIN = 4.2V TA = 25C
9
10
50
20
30
40
50 60 70 80 DUTY CYCLE (%)
90
100
3700 G10
3700 G05
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LTC3700 TYPICAL PERFOR A CE CHARACTERISTICS
LDO REGULATOR
VFB2 Voltage vs Temperature
850 840 830 VFB2 VOLTAGE (mV) VFB2 VOLTAGE (mV) 820 810 800 790 780 770 760 750 -55 -35 -15 5 25 45 65 85 105 125 TEMPERATURE (C)
3700 G06
VIN2 = 4.2V LDO = VFB2 ILOAD = 1mA ILOAD = 10A
LDO Pass FET RON vs Input Voltage
4.0 3.7 3.4 3.1 VIN = 0 ILDO = 100mA TA = 25C
PGOOD RON ()
RON ()
2.8 2.5 2.2 1.9 1.6 1.3 1.0 2 2.5 3 3.5 4 4.5 5 VIN2 INPUT VOLTAGE (V) 5.5 6
LDO Supply Current vs Input Voltage
LDO = VFB2 110 ILDO = 10A T = 25C 100 A 90 80 70 60 50 40 30 20 2 2.5 3.5 4 3 4.5 5 VIN2 INPUT VOLTAGE (V) 5.5 6 VIN = 9.8V 120
VIN = 0V
VIN2 SUPPLY CURRENT (A)
UW
LDO Line Regulation (VFB2 Voltage vs Supply)
850 840 830 820 810 800 790 780 770 760 750 2.4 2.85 3.3 3.75 4.2 4.65 5.1 5.55 VIN2 INPUT VOLTAGE (V) 6 ILOAD = 10mA ILOAD = 100mA ILOAD = 10A ILOAD = 1mA TA = 25C LDO = VFB2
ILOAD = 10mA ILOAD = 100mA
3700 G07
PGOOD RON vs Input Voltage
300 270 240 210 180 150 120 90 60 30 0 2 3 4 5 6 7 8 VIN INPUT VOLTAGE (V) 9 10 VIN2 = 0V VPGOOD = 100mV TA = 25C
3700 G08
3700 G09
Load Transient Response
150 ILDO (mA) 100 50mA/DIV 50 0 VLDO 20mV/DIV AC COUPLED
0
TA = 25C VIN2 = 3.3V VLDO = 2.5V CLDO = 10F
20s/DIV
3700 G12
3700 G11
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LTC3700
PIN FUNCTIONS
VIN2 (Pin 1): LDO Input Supply Pin. Must be closely decoupled to GND (Pin 5). LDO (Pin 2): LDO Output Pin. Must be closely decoupled to GND (Pin 5) with a low ESR ceramic capacitor 2.2F. VFB2 (Pin 3): LDO Feedback Voltage. Receives the feedback voltage from an external resistor divider between LDO (Pin 2) and GND (Pin 5). PGOOD (Pin 4): Open-Drain Power Good Output. This pin will pull to ground if either voltage output of the buck or the LDO [sensed at VFB (Pin 9) and VFB2 (Pin 3), respectively] is out of range. When both voltage outputs are valid, this pin will go to a high impedance state. GND (Pin 5): Common Ground Pin for Both Buck and LDO. PGATE (Pin 6): Gate Drive for Buck's External P-Channel MOSFET. This pin swings from 0V to VIN. VIN (Pin 7): Buck Input Supply Pin. Must be closely decoupled to GND (Pin 5). SENSE - (Pin 8): The Negative Input to the Current Comparator of the Buck. Monitors switch current of external P-Channel MOSFET. VFB (Pin 9): Buck Feedback Voltage. Receives the feedback voltage from an external resistor divider between buck output and GND (Pin 5). ITH/RUN (Pin 10): This pin performs two functions. It serves as the error amplifier compensation point for the buck, as well as a common run control input for both the buck and the LDO. The current comparator threshold of the buck increases with this voltage. Nominal voltage range for this pin is 0.7V to 1.9V. Forcing this pin below 0.3V causes both the buck and the LDO to be shut down. In shutdown all functions are disabled, the PGATE pin is held high and the LDO output will go to a high impedance state.
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LTC3700 FUNCTIONAL DIAGRA W
PGOOD 4 0.86V VFB VFB2 0.74V PGOOD 0.8V VFB2 3 4 VIN2 1
VIN 7
FREQ FOLDBACK SHORT-CIRCUIT DETECT
VIN 0.5A
+
VIN 0.3V VIN2
-
VOLTAGE REFERENCE GND 5 UNDERVOLTAGE LOCKOUT VREF 0.8V
-
+
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SENSE - 8 4
-
LDO
LDO 2
+
SHDN OVERTEMPERATURE DETECT
+
ICMP
-
RS1 OSC SLOPE COMP R Q S SWITCHING LOGIC AND BLANKING CIRCUIT VIN PGATE 6
0.3V 0.15V
+ -
BURST CMP
OVP
+ -
VREF + 60mV
SLEEP
EAMP
+ -
VREF 0.8V VFB 9 VIN
10 ITH/RUN
0.3V
+
SHDN CMP SHDN UV
-
1.2V
3700 FD
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LTC3700
OPERATIO
Main Control Loop (Buck Controller) The LTC3700 is a constant frequency current mode switching regulator. During normal operation, the external P-channel power MOSFET is turned on each cycle when the oscillator sets the RS latch (RS1) and turned off when the current comparator (ICMP) resets the latch. The peak inductor current at which ICMP resets the RS latch is controlled by the voltage on the ITH/RUN pin, which is the output of the error amplifier EAMP. An external resistive divider connected between VOUT and ground allows the EAMP to receive an output feedback voltage VFB. When the load current increases, it causes a slight decrease in VFB relative to the 0.8V reference, which in turn causes the ITH/RUN voltage to increase until the average inductor current matches the new load current. The main control loop is shut down by pulling the ITH/RUN pin low. Releasing ITH/RUN allows an internal 0.5A current source to charge up the external compensation network. When the ITH/RUN pin reaches 0.3V, the main control loop is enabled with the ITH/RUN voltage then pulled up to its zero current level of approximately 0.7V. As the external compensation network continues to charge up, the corresponding output current trip level follows, allowing normal operation. Comparator OVP guards against transient overshoots > 7.5% by turning off the external P-channel power MOSFET and keeping it off until the fault is removed. Burst Mode Operation The buck enters Burst Mode operation at low load currents. In this mode, the peak current of the inductor is set as if VITH/RUN = 1V (at low duty cycles) even though the voltage at the ITH/RUN pin is at a lower value. If the inductor's average current is greater than the load requirement, the voltage at the ITH/RUN pin will drop. When the ITH/RUN voltage goes below 0.85V, the sleep signal goes high, turning off the external MOSFET. The sleep signal goes low when the ITH/RUN voltage goes above 0.925V and the buck resumes normal operation. The next oscillator cycle will turn the external MOSFET on and the switching cycle repeats. Overvoltage Protection As a further protection, the overvoltage comparator in the buck will turn the external MOSFET off when the feedback voltage has risen 7.5% above the reference voltage of 0.8V. This comparator has a typical hysteresis of 20mV. Slope Compensation and Inductor's Peak Current The inductor's peak current is determined by:
IPK = VITH - 0.7 10(RSENSE )
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(Refer to Functional Diagram)
Dropout Operation When the input supply voltage decreases towards the output voltage, the rate of change of inductor current during the ON cycle decreases. This reduction means that the external P-channel MOSFET will remain on for more than one oscillator cycle since the inductor current has not ramped up to the threshold set by EAMP. Further reduction in input supply voltage will eventually cause the P-channel MOSFET to be turned on 100%, i.e., DC. The output voltage will then be determined by the input voltage minus the voltage drop across the MOSFET, the sense resistor and the inductor. Undervoltage Lockout To prevent operation of the P-channel MOSFET below safe input voltage levels, an undervoltage lockout is incorporated into the buck input supply. When the input supply voltage drops below approximately 2.1V, the P-channel MOSFET and all circuitry is turned off except the undervoltage block, which draws only several microamperes. Short-Circuit Protection When the output is shorted to ground, the frequency of the oscillator will be reduced to about 110kHz. This lower frequency allows the inductor current to safely discharge, thereby preventing current runaway. The oscillator's frequency will gradually increase to its designed rate when the feedback voltage again approaches 0.8V.
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LTC3700
OPERATIO
when the buck is operating below 40% duty cycle. However, once the duty cycle exceeds 40%, slope compensation begins and effectively reduces the peak inductor current. The amount of reduction is given by the curves in Figure 2. Soft-Start An internal default soft-start circuit is employed at power up and/or when coming out of shutdown. The soft-start circuit works by internally clamping the voltage at the ITH/ RUN pin to the corresponding zero-current level and gradually raising the clamp voltage such that the minimum time required for the programmed switch current to reach its maximum is approximately 0.5msec. After the softstart circuit has timed out, it is disabled until the part is put in shutdown again or the input supply is cycled. LDO Regulator The 150mA low dropout (LDO) regulator on the LTC3700 employs an internal P-channel MOSFET pass device between its input supply (VIN2) and the LDO output pin. The pass FET has an on-resistance of approximately 1.5 (with VIN2 = 4.2V) with a strong dependence on input supply voltage. The dropout voltage is simply the FET onresistance multiplied by the load current when in dropout.
110 100 90
SF = IOUT/IOUT(MAX) (%)
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(Refer to Functional Diagram)
The LDO is protected by both current limit and thermal shutdown circuits. Current limit is set such that the output voltage will start dropping out when the load current reaches approximately 200mA. With a short-circuited LDO output, the device will limit the sourced current to approximately 225mA. The thermal shutdown circuit has a typical trip point of 150C with a typical hysteresis of 5C. In thermal shutdown, the LDO pass device is turned off. Frequency compensation of the LDO is accomplished by forcing the dominant pole at the output. For stability, a low ESR ceramic capacitor 2.2F is required from LDO to GND. For improved transient response, particularly at heavy loads, it is recommended to use the largest value of capacitor available in the same size considered. Both the buck and the LDO share the same internally generated bandgap reference voltage for their feedback reference. When both input supplies are present, the internal reference is powered by the buck input supply (VIN). For this reason, line regulation for the LDO output is specified both with respect to VIN and VIN2 if the buck is present and with respect only to VIN2 if the buck is disabled. The same is true for VIN2 supply current, which will be higher when the buck is disabled by the current draw of the internal reference.
80 70 60 50 40 30 20 10 0 10 20 30 40 50 60 70 80 90 100 DUTY CYCLE (%)
3700 F02
IRIPPLE = 0.4IPK AT 5% DUTY CYCLE IRIPPLE = 0.2IPK AT 5% DUTY CYCLE VIN = 4.2V
Figure 2. Maximum Output Current vs Duty Cycle
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LTC3700
APPLICATIONS INFORMATION
The basic LTC3700 application circuit is shown in Figure 1. External component selection for the buck is driven by the load requirement and begins with the selection of L1 and RSENSE (= R1). Next, the power MOSFET, M1 and the output diode D1 are selected followed by CIN (= C1) and COUT (= C2). RSENSE Selection for Output Current RSENSE is chosen based on the required output current. With the current comparator monitoring the voltage developed across RSENSE, the threshold of the comparator determines the inductor's peak current. The output current the buck can provide is given by:
IOUT =
0.12 I - RIPPLE RSENSE 2
where IRIPPLE is the inductor peak-to-peak ripple current (see Inductor Value Calculation section). A reasonable starting point for setting ripple current is IRIPPLE = (0.4)(IOUT). Rearranging the above equation, it becomes:
RSENSE = 1 for Duty Cycle < 40% (10)(IOUT )
However, for operation that is above 40% duty cycle, slope compensation effect has to be taken into consideration to select the appropriate value to provide the required amount of current. Using Figure 2, the value of RSENSE is:
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RSENSE =
SF (10)(IOUT )(100)
Inductor Value Calculation The operating frequency and inductor selection are interrelated in that higher operating frequencies permit the use of a smaller inductor for the same amount of inductor ripple current. However, this is at the expense of efficiency due to an increase in MOSFET gate charge losses. The inductance value also has a direct effect on ripple current. The ripple current, IRIPPLE, decreases with higher inductance or frequency and increases with higher VIN or VOUT. The inductor's peak-to-peak ripple current is given by:
IRIPPLE =
VIN - VOUT VOUT + VD f(L) VIN + VD
where f is the operating frequency. Accepting larger values of IRIPPLE allows the use of low inductances, but results in higher output voltage ripple and greater core losses. A reasonable starting point for setting ripple current is IRIPPLE = 0.4(IOUT(MAX)). Remember, the maximum IRIPPLE occurs at the maximum input voltage.
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LTC3700
APPLICATIONS INFORMATION
In Burst Mode operation on the LTC3700, the ripple current is normally set such that the inductor current is continuous during the burst periods. Therefore, the peakto-peak ripple current must not exceed: Molypermalloy (from Magnetics, Inc.) is a very good, low loss core material for toroids, but it is more expensive than ferrite. A reasonable compromise from the same manufacturer is Kool M. Toroids are very space efficient, especially when you can use several layers of wire. Because they generally lack a bobbin, mounting is more difficult. However, new designs for surface mount that do not increase the height significantly are available. Power MOSFET Selection An external P-channel power MOSFET must be selected for use with the LTC3700. The main selection criteria for the power MOSFET are the threshold voltage VGS(TH) and the "on" resistance RDS(ON), reverse transfer capacitance CRSS and total gate charge. Since the LTC3700 is designed for operation down to low input voltages, a sublogic level threshold MOSFET (RDS(ON) guaranteed at VGS = 2.5V) is required for applications that work close to this voltage. When these MOSFETs are used, make sure that the input supply to the buck is less than the absolute maximum VGS rating, typically 8V. The required minimum RDS(ON) of the MOSFET is governed by its allowable power dissipation. For applications that may operate the LTC3700 in dropout, i.e., 100% duty cycle, at its worst case the required RDS(ON) is given by: R DS(ON)
DC=100%
IRIPPLE
0.03 RSENSE
This implies a minimum inductance of: V + VD V -V LMIN = IN OUT OUT 0.03 VIN + VD f RSENSE (Use VIN(MAX) = VIN) A smaller value than L MIN could be used in the circuit; however, the inductor current will not be continuous during burst periods. Inductor Core Selection Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy or Kool M(R) cores. Actual core loss is independent of core size for a fixed inductor value, but it is very dependent on inductance selected. As inductance increases, core losses go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core losses and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates "hard," which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent output voltage ripple. Do not allow the core to saturate!
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=
(IOUT(MAX) )2 (1+ p)
PP
where PP is the allowable power dissipation and p is the temperature dependency of RDS(ON). (1 + p) is generally given for a MOSFET in the form of a normalized RDS(ON) vs temperature curve, but p = 0.005/C can be used as an approximation for low voltage MOSFETs.
Kool M is a registered trademark of Magnetics, Inc.
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LTC3700
APPLICATIONS INFORMATION
In applications where the maximum duty cycle is less than 100% and the buck is in continuous mode, the RDS(ON) is governed by: RDS(ON) where PD is the allowable power dissipation and will be determined by efficiency and/or thermal requirements. A fast switching diode must also be used to optimize efficiency. Schottky diodes are a good choice for low forward drop and fast switching times. Remember to keep lead length short and observe proper grounding (see Board Layout Checklist) to avoid ringing and increased dissipation. CIN and COUT Selection In continuous mode, the source current of the P-channel MOSFET is a square wave of duty cycle (VOUT + VD)/ (VIN + VD). To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by:
(DC )IOUT 2 (1+ p)
PP
where DC is the maximum operating duty cycle of the buck. Output Diode Selection The catch diode carries load current during the off-time. The average diode current is therefore dependent on the P-channel switch duty cycle. At high input voltages the diode conducts most of the time. As VIN approaches VOUT the diode conducts only a small fraction of the time. The most stressful condition for the diode is when the output is short-circuited. Under this condition the diode must safely handle IPEAK at close to 100% duty cycle. Therefore, it is important to adequately specify the diode peak current and average power dissipation so as not to exceed the diode ratings. Under normal load conditions, the average current conducted by the diode is:
V -V ID = IN OUT IOUT VIN + VD
The allowable forward voltage drop in the diode is calculated from the maximum short-circuit current as:
VF PD ISC(MAX)
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[VOUT (VIN - VOUT )]1/ 2 CIN Required IRMS IMAX
VIN This formula has a maximum value at VIN = 2VOUT, where IRMS = IOUT /2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturer's ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. Several capacitors may be paralleled to meet the size or height requirements in the design. Due to the high operating frequency of the LTC3700, ceramic capacitors can also be used for CIN. Always consult the manufacturer if there is any question.
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LTC3700
APPLICATIONS INFORMATION
The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering. The output ripple (VOUT) is approximated by: In surface mount applications, multiple capacitors may have to be paralleled to meet the ESR or RMS current handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS, AVX TPSV and KEMET T510 series of surface mount tantalum, available in case heights ranging from 2mm to 4mm. Other capacitor types include Sanyo OS-CON, Nichicon PL series and Panasonic SP. Low Supply Voltage Operation Although the LTC3700 can function down to 2.1V (typ), the maximum allowable output current is reduced when VIN decreases below 3V. Figure 3 shows the amount of change as the supply is reduced down to 2.2V. Also shown in Figure 3 is the effect of VIN on VREF as VIN goes below 2.3V.
1 VOUT IRIPPLE ESR + 8 fCOUT
where f is the operating frequency, COUT is the output capacitance and IRIPPLE is the ripple current in the inductor. The output ripple is highest at maximum input voltage since IL increases with input voltage. Manufacturers such as Nichicon, United Chemicon and Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest ESR (size) product of any aluminum electrolytic at a somewhat higher price. Once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement.
105 100 95 90 85 80 75 2.0 VREF
NORMALIZED VOLTAGE (%)
Figure 3. Line Regulation of VREF and VITH
U
W
U
U
VITH
2.2
2.4 2.6 2.8 INPUT VOLTAGE (V)
3.0
3700 F03
3700f
13
LTC3700
APPLICATIONS INFORMATION
Setting Output Voltage (Buck Controller) The buck develops a 0.8V reference voltage between the feedback (Pin 9) terminal and ground (see Figure 4). By selecting resistor R1, a constant current is caused to flow through R1 and R2 to set the overall output voltage. The regulated output voltage is determined by: R2 VOUT1 = 0.8 1 + R1 For most applications, an 80k resistor is suggested for R1. To prevent stray pickup, locate resistors R1 and R2 close to LTC3700. Foldback Current Limiting As described in the Output Diode Selection, the worstcase dissipation occurs with a short-circuited output when the diode conducts the current limit value almost continuously. To prevent excessive heating in the diode, foldback current limiting can be added to reduce the current in proportion to the severity of the fault. Foldback current limiting is implemented by adding diodes DFB1 and DFB2 between the output and the ITH/RUN pin as shown in Figure 5. In a hard short (VOUT = 0V), the current will be reduced to approximately 50% of the maximum output current. Setting Output Voltage (LDO Regulator) The LDO develops a 0.8V reference voltage between VFB2 (Pin 3) and ground (see Figure 6), similar to the buck controller. The regulated output voltage VOUT2 is given by: R4 VOUT2 = 0.8 1 + R3 For most applications, an 80k resistor is suggested for R3. To prevent stray pickup, locate resistors R3 and R4 close to LTC3700.
VOUT1 LTC3700 VFB R2 9
R1
3700 F04
Figure 4. Setting Output Voltage (Buck Controller)
LTC3700 VFB2 3
Figure 6. Setting Output Voltage (LDO Regulator)
14
U
W
U
U
LTC3700 R2 10 ITH /RUN VFB 9 R1 DFB2
3700 F05
VOUT1
+
DFB1
Figure 5. Foldback Current Limiting
LDO
2 R4
VOUT2
R3
3700 F06
3700f
LTC3700
PACKAGE DESCRIPTION
MS Package 10-Lead Plastic MSOP
(Reference LTC DWG # 05-08-1661)
5.23 (.206) MIN
0.50 0.305 0.038 (.0197) (.0120 .0015) BSC TYP RECOMMENDED SOLDER PAD LAYOUT
0.254 (.010) GAUGE PLANE
0.18 (.007) SEATING PLANE 0.17 - 0.27 (.007 - .011) TYP 0.13 0.076 (.005 .003)
MSOP (MS) 0802
NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
U
0.889 0.127 (.035 .005)
3.2 - 3.45 (.126 - .136) 3.00 0.102 (.118 .004) (NOTE 3) 10 9 8 7 6
0.497 0.076 (.0196 .003) REF
DETAIL "A" 0 - 6 TYP
4.90 0.15 (1.93 .006)
3.00 0.102 (.118 .004) NOTE 4
12345 0.53 0.01 (.021 .006) DETAIL "A" 1.10 (.043) MAX 0.86 (.034) REF
0.50 (.0197) BSC
3700f
15
LTC3700
TYPICAL APPLICATIO
VIN1 5V
5V Input Supply to 3.3V/1A High Efficiency Output and 2.5V/150mA Low Noise Output
VOUT1 3.3V AT 1A C2 47F 6V
+
RELATED PARTS
PART NUMBER LT1375/LT1376 LTC1622 LTC1624 LTC1625 LTC1649 LTC1702A LTC1704 LTC1735 LT1761 LTC1771 LTC1772 LTC1773 LTC1778 LTC1872 LTC3404 DESCRIPTION 1.5A, 500kHz Step-Down Switching Regulators Low Input Voltage Current Mode Step-Down DC/DC Controller High Efficiency SO-8 N-Channel Switching Regulator Controller No RSENSE Synchronous Step-Down Regulator 3.3V Input Synchronous Controller 550kHz, 2 Phase, Dual Synchronous Controller
TM
Synchronous Step-Down Controller Plus Linear Regulator Controller No Current Sense Required, N-Channel MOSFET Drivers, Adjustable Soft-Start Single, High Efficiency, Low Noise Synchronous Switching Controller High Efficiency 5V to 3.3V Conversion at up to 15A 100mA, Low Noise, LDO Micropower Regulators in ThinSOTTM Ultralow Supply Current Step-Down DC/DC Controller Constant Frequency Current Mode Step-Down DC/DC Controller in ThinSOT 95% Efficient Synchronous Step-Down Controller No RSENSE Current Mode Synchronous Step-Down Controller ThinSOT Step-Up Controller 1.4MHz Monolithic Synchronous Step-Down Controller 1.8V VIN 20V, 300mV Dropout at 100mA 10A Supply Current, 93% Efficiency, 1.23V VOUT 18V, 2.8V VIN 20V With Burst Mode Operation for Higher Efficiency at Light Load Current 2.65V VIN 8.5V, 0.8V VOUT VIN, Current Mode, 550kHz Up to 97% Efficiency, 4V VIN 36V, 0.8V VOUT (0.9)(VIN), IOUT up to 20A 2.5V VIN 9.8V, 550kHz, 90% Efficiency 2.65V VIN 6V, 700mA Output Current, 8-Lead MSOP VIN = 2.5V to 5.5V, 95% Efficiency, ThinSOT, B Version: Burst Mode Defeat
LTC3406/LTC3406B 600mA (IOUT), 1.5MHz Synchronous Step-Down Converter No RSENSE and ThinSOT are trademarks of Linear Technology Corporation.
16
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 q FAX: (408) 434-0507
q
U
C1 10F 16V L1 15H 249k M1 R1 0.05 8 6 7 VIN SENSE - PGATE VFB2 D1 LTC3700 C1: TAIYO YUDEN EMK325BJ106MNT C2: SANYO POSCAP 6TPA47M C3: MURATA GRM42-6X7R225K016AL D1: MOTOROLA MBRS130LT3 L1: COILTRONICS UP1B150 M1: Si3443DV 3700 TA01 R1: DALE 0.25W 1 VIN2 LDO 2 169k 3 78.7k VOUT2 2.5V AT 150mA C3 2.2F 16V 80.6k 9 10 220pF 10k VFB PGOOD GND 4 5 ITH/RUN
COMMENTS High Frequency, Small Inductor, High Efficiency VIN 2V to 10V, IOUT Up to 4.5A, Synchronizable to 750kHz Optional Burst Mode Operation, 8-Lead MSOP N-Channel Drive, 3.5V VIN 36V 97% Efficiency, No Sense Resistor, Current Mode No Need for 5V Supply, Uses Standard Logic Gate MOSFETs, IOUT up to 15A Two Channels, Minimum CIN and COUT, IOUT up to 15A
3700f LT/TP 0203 2K * PRINTED IN USA
www.linear.com
(c) LINEAR TECHNOLOGY CORPORATION 2001

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